With the trend of power supplies for portable systems continuously decreasing to lower supply voltages, analog designs must be adjusted proportionately to be operable within the lower supply voltage requirement. Of all analog circuit designs, operational amplifiers are a challenge to modify in that an operational amplifier typically requires high open-loop gain and high-frequency response to minimize errors in the output voltage. In addition, operational amplifiers typically require high output swing to maximize the signal-to-noise ratio, especially in low-supply applications. Obtaining high open-loop gain may be achieved by increasing the output resistance. One way to increase output resistance is through the use of a folded cascode. In addition, gain boosting of the cascode devices may enhance the open-loop gain. An increase in output resistance, however, exists often at the expense of output swing. Thereby, in amplifiers that employ gain-boosting of cascode devices to achieve high gain, there exists a problem in achieving a wide output swing near the power supplies without significant complexity or distortion.
Specifically, a simplified version of a cascode (not shown) comprises a top and bottom cascode transistor connected such that the drain of the top couples to the source of the bottom cascode transistor. This simple cascode increases the output resistance by a factor of gmro, wherein gm is the small-signal transconductance and ro is the small signal output resistance. Thereby, the voltage gain is increased by the same factor. Although a voltage equivalent to the drain-to-source saturation voltage VDS,SAT is necessary to saturate the bottom cascoded transistor, a safety margin voltage Vmargin is added to ensure that the bottom transistor operates in the saturation region. Thus, taking into account the voltage necessary to keep both cascoded transistors in saturation, the maximum swing from supply is two times the drain-to-source saturation voltage VDS,SAT plus the margin voltage Vmargin. Since the saturation voltage VDS,SAT and the margin voltage Vmargin are approximately 200 mV, the difference in output swing can be very large with respect to a low power supply. The addition of a gain boosting amplifier increases the output resistance by the additional gain of A, where A is the gain of the amplifier. The output swing limitation, however, further increases by one threshold voltage VT, wherein the output swings to within twice the saturation voltage VDS,SAT, the margin voltage Vmargin, and one threshold voltage VT.
Specifically, referring to FIG. 1, amplifiers A1, A2, A3, and A4 amplify the gate inputs of transistors, MP3, MP4, MN3, and MN4, respectively. Amplifiers A1, A2, A3 and A4, provide an increase in the output resistance through boosting the gain of the cascode devices MP3, MP4, MN3 and MN4. Accordingly, the output resistance is increased as is shown in the following equation:
      r    out    ≈      1                  g        ds1            /                                    (                          A              +              1                        )                    ⁢                      g            m3                                    g          ds3                    wherein gm3 is the small-signal transconductance; and gds1 and gds3 are the transconductance relative to the drain-to-source connection of transistors MP1 and MP3, respectively. Amplifiers A1, A2, A3 and A4, are added in an effort to achieve a high output impedance, wherein the output nodes swing very close to the supply rails. Transistors MP3, MP4, MN3 and MN4, however, shift out of saturation and into the triode region, when these transistors MP3, MP4, MN3 and MN4 should remain in saturation. Initially, when transistors, MP3, MP4, MN3 and MN4, shift into the triode region, the gain and the large output resistance at each output node is lost. In an effort to regain the large output resistance and thereby increase the gain of the operational amplifier, there is a need for a cascode circuit design that keeps transistors, MP1–MP4 and MN1–MN4, biased in saturation.
More particularly, in FIG. 1, amplifier A1 boosts the small-signal transconductance gm3 of transistor MP3. Specifically, the source of transistor MP3 couples to an input of amplifier A1 and amplifier A1 couples to receive the bias voltage Vbias1. The feedback from the source of transistor MP3 guarantees that the source of transistor MP3 will always be equal to voltage Vbias1. The objective is for the drain-to-source voltages of transistors, MP1 and MP3, to be small and always in saturation. As a result, the output resistance is multiplied by the value of the amplifier A plus one. The objective as explained previously is to bias each cascode transistor such that the outputs, 30 and 32, of the cascode stage are enabled to swing close to either power supply rail. The common-mode feedback circuit 34 controls the common-mode of these differential outputs, 30 and 32.
FIGS. 2A–D display various known simple amplifier designs, 40, 42, 44, and 46. Specifically, the amplifiers, 40 and 42, of FIGS. 2A and 2B include a current mirror transistor pair, MP20, MP21, MN22, and MN23, respectively. Input transistors MN20 and MN21 of amplifier 40 connect to the respective legs of the current mirror transistor pair, MP20 and MP21. A differential input, IN1 and IN2, couples across the gates of transistors, MN20 and MN21. Accordingly, an output terminal Out is formed by the drain of both FETs, MP21 and MN21. A current source I1 couples to the source nodes of FETs, MN20 and MN21. FIG. 2B illustrates the p-type amplifier 42, wherein the differential input, IN3 and IN4, couples across the gates of transistors, MP22 and MP23. A current source 12 connects to the source of each transistor, MP22 and MP23, wherein each transistor couples to a respective leg of the current mirror, MN22, and MN23.
In the alternative, known amplifier designs, 44 and 46, include current mirrors made from respective transistor pairs, MN24, MN25, MP24 and MP25. Specifically, the current mirror, MN24 and MN25, connects to separate respective inputs instead of connecting to input transistors as is shown in FIGS. 2A and 2B. Current sources I3 and I4 of amplifier 44 connect the current mirror, MN24 and MN25. FIG. 2D illustrates the n-type transistor amplifier 44 version of the amplifier 46 of FIG. 2C.
Since the amplifier A1 of FIG. 1 must supply voltage to the gate and source of transistor MP3 and the drain of transistor MP1, the source-to-drain voltages of both transistors, MP1 and MP3, must be extremely small to enable that the drain of MP3 to swing close to the supply rail voltage. At the maximum, the drain-to-source voltages of either transistor, MP1 and MP3, may be 400 mV. As a result the inputs of amplifier A1 will be within the range of a few millivolts (mV) away from the supply rail voltage. If, however, each input, IN1 and IN2, of amplifier 40, for example, are 400 mV below the supply rail voltage, there will not be enough headroom for the p-channel mirror, MP20 and MP21.
One approach to generate an increase in the voltage headroom at the drains of transistors, MP26 and MP27, is to couple additional transistors MN28, MN29, MN30, and MN31, configured as source followers at the inputs or gates of MN26 and MN27 of amplifier 50 as is shown in FIG. 3. These additional transistors configured as source followers will provide level shifting of the voltage at the drain nodes of each transistor of the current mirror, MP26 and MP27. The objective is to enable the inputs, IN9 and IN10, to be very close to both supply rails. This design, however, is not an efficient one since an additional number of transistors are needed to effectively level shift the voltage. Disadvantageously, these additional transistors do not contribute to the gain of the boosting amplifier.
Some other solutions employ more complex gain-boosting amplifiers for obtaining a wide-swing for the cascode stage. These designs are not efficient, however, due to their complexity. Another simple alternative includes connecting the source of a transistor to the source of the cascode input transistor. This solution, however, can increase distortion when the cascode is used in the signal path such as a folded cascode amplifier.
Thus, there still exists a need for an gain-boosting amplifier design that provides level shifting of the voltages without much complexity. Particularly, this type of implementation is essential in the case where there is a need for high bandwidth within the amplifier design.
The present invention is directed to overcoming, or at least reducing the effects of one or more of the problems set forth above.